The Advanced Television Systems Committee (ATSC) published a Digital Television Standard in 1995 as Document A/53, hereinafter referred to simply as “A/53” for sake of brevity. Annex D of A/53 titled “RF/Transmission Systems Characteristics” is particularly incorporated by reference into this specification. Annex D specifies that the data frame shall be composed of two data fields, each data field composed of 313 data segments, and each data segment composed of 832 symbols. The 8-level symbols in the 8VSB signal have normalized modulation values of −7, −5, −3, −1, +1, +3, +5 and +7 superposed on a +1.25 normalized modulation value associated with the production of a pilot carrier signal. The 8-level symbols result from ⅔ trellis coding of successive bit pairs or “nibbles”. Each 8-level symbol codes a Z2 most significant bit, a Z1 next most significant bit, and a Z0 least significant bit derived from preceding Z1 bits. Annex D specifies that each data segment shall begin with a 4-symbol data-segment-synchronization (DSS) sequence rendered as successive +5, −5, −5 and +5 normalized modulation values. Annex D specifies that the initial data segment of each data field shall contain a data-field synchronization (DFS) signal following the 4-symbol DSS sequence therein. The fifth through 515th symbols in each A/53 DFS signal are a specified PN511 sequence—that is a pseudo-random noise sequence composed of 511 symbols which are rendered as +5 or −5 normalized modulation values. The 516th through 704th symbols in each A/53 DFS signal are a triple-PN63 sequence composed of a total of 189 symbols which are rendered as +5 or −5 normalized modulation values. The middle PN63 sequence is inverted in polarity every other data field. The 705th through 728th symbols in each A/53 DFS signal contain a VSB mode code specifying the nature of the vestigial-sideband (VSB) signal being transmitted. The remaining 104 symbols in the each A/53 DFS signal were reserved, with the last twelve of these symbols being a precode signal that repeats the last twelve symbols of the data in the last data segment of the previous data field. A/53 specifies such precode signal to implement trellis coding and decoding procedures being able to resume in the second data segment of each field proceeding from where those procedures left off processing the data in the preceding data field.
The broadcast TV signal to which the receiver synchronizes its operations is called the principal signal, and the principal signal is usually the direct signal received over the shortest transmission path. Thus, the multipath signals received over other paths are usually delayed with respect to the principal signal and appear as lagging ghost signals. It is possible however, that the direct or shortest path signal is not the signal to which the receiver synchronizes. When the receiver synchronizes its operations to a (longer path) signal that is delayed respective to the direct signal, there will be a leading multipath signal caused by the direct signal. Often there will a plurality of leading multipath signals, caused by the direct signal and other reflected signals of lesser delay than the reflected signal to which the receiver synchronizes. Multipath signals are referred to as “ghosts” in the analog TV art, but in the DTV art multipath signals are customarily referred to as “echoes”. The multipath signals that lead the principal signal are referred to as “pre-echoes”, and the multipath signals that lag the principal signal are referred to as “post-echoes”. The echoes vary in number, amplitude and delay time from location to location and from channel to channel at a given location. Post-echoes with significant energy have been reported as being delayed from the reference signal by as many as sixty microseconds. Pre-echoes with significant energy have been reported leading the reference signal by as many as thirty microseconds. This 90-microsecond or so possible range of echoes of is appreciably more extensive than was generally supposed before spring 2000.
The transmission of DTV signal to the receiver is considered to be through a transmission channel that has the characteristics of a sampled-data time-domain filter that provides weighted summation of variously delayed responses to the transmitted signal. In the DTV signal receiver the received signal is passed through equalization and echo-cancellation filtering that compensates at least partially for the time-domain filtering effects that originate in the transmission channel. This equalization and echo-cancellation filtering is customarily sampled-data filtering performed in the digital domain. Time-domain filtering effects differ for the channels through which broadcast digital television signals are received from various transmitters. Furthermore, time-domain filtering effects change over time for the broadcast digital television signals received from each particular transmitter. Changes referred to as “dynamic multipath” are introduced while receiving radio-frequency signal from a single transmitter when the lengths of reflective transmission paths change, owing to the reflections being from moving objects. Accordingly, adaptive filter procedures are required for adjusting the weighting coefficients of the sampled-data filtering that provides echo-cancellation and equalization.
Computation of the weighting coefficients of the sampled-data filtering that provides equalization and echo-cancellation is customarily attempted using one or more methods of three distinctive general types. Methods of the Wiener type adjust the weighting coefficients of the sampled-data filtering incrementally using auto-regressive decision-feedback based on the effects of multipath just on intermittent echo-cancellation reference (ECR) signals included in the transmitted signal specifically to facilitate such analysis. Methods of the Kalman type adjust the weighting coefficients of the sampled-data filtering incrementally using auto-regressive decision-feedback based on the effects of multipath on all portions of the transmitted signal. Methods of the Dietrich-Greenberg type calculate the weighting coefficients by DFT methods based on the effects of multipath on ECR signals that are repetitive PN sequences. See U.S. Pat. No. 5,065,242 titled “Deghosting apparatus using pseudorandom sequences”, which was granted 23 Aug. 1994 to C. B. Dietrich and A. Greenberg.
While the PN511 and triple-PN63 sequences in the initial data segments of the data fields in the ATSC standard DTV signal were originally proposed for use as ECR signals to implement the computation of the weighting coefficients of the sampled-data filtering by a method of Dietrich-Greenberg type, the VSB receiver performance in actual field environments has demonstrated that these sequences inadequately support such computation. So, most DTV manufacturers have used decision-feedback methods of Kalman type that rely on analysis of the effects of multipath on all portions of the transmitted signal for adapting the weighting coefficients of the sampled-data filtering. Decision-feedback methods that utilize least-mean-squares (LMS) method or block LMS method can be implemented in an integrated circuit of reasonable size. Kalman-type decision-feedback provides for tracking dynamic multipath conditions reasonably well after the equalization and echo-cancellation filtering has initially been converged to substantially optimal response by Wiener-type decision-feedback. That is, providing that the sampling rate through the filtering is appreciably higher than symbol rate and providing that the rate of change of the dynamic multipath does not exceed the slewing rate of the decision-feedback loop.
However, these decision-feedback methods tend to be unacceptably slow in converging the equalization and echo-cancellation filtering to nearly optimal response when initially receiving a DTV signal that has severe multipath distortion. Severe multipath distortion conditions include cases where echoes of substantial energy lead or lag the principal received signal by more than ten or twenty microseconds, cases where there is an ensemble of many echoes with differing timings relative to the principal received signal, cases where multipath distortion changes rapidly, and cases where it is difficult to distinguish principal received signal from echo(es) because of similarity in energy level.
Worse yet, convergence is too slow when tracking of dynamic multipath conditions must be regained after the slewing rate of the decision-feedback loop has not been fast enough to keep up with rapid change in the multipath conditions. Data dependent equalization and echo cancellation methods that provide faster convergence than LMS or block-LMS decision-feedback methods are known, but there is difficulty in implementing them in an integrated circuit of reasonable size.
Accordingly, it is desirable to modify A/53 DTV signal to introduce periodically an ECR signal that will “instantly” converge the equalization and echo-cancellation filtering to substantially optimally response. A repetitive pseudo-random noise (PN) sequence has uniquely strong auto-correlation properties, particularly when wrapped around to fit a cylindrical coordinate system. An auto-correlation response corresponding to the impulse response of the channel supplying a repeating long PN sequence is simply generated by passing an interior cycle of the repeating sequence through a convolution filter with a kernel having weights in accordance with the PN sequence. U.S. Pat. No. 6,816,204 granted to A. L. R. Limberg on 9 Nov. 2004 and titled “Ghost cancellation reference signals for broadcast digital television signal receivers and receivers for utilizing them” describes the insertion of repetitive PN511 sequences into 8VSB digital television signals such that data segment sync signals are subsumed into the sequences. Limberg presumed an echo range of only 45 microseconds or so, and the ECR signals specifically described rely on repetitive PN511 sequences with baud-rate symbols rendered as +5 or −5 values. U.S. Pat. No. 6,768,517 granted to A. L. R. Limberg, J. D. McDonald and C. B. Patel on 27 Jul. 2004 and titled “Repetitive-PN1023-sequence echo-cancellation reference signal for single-carrier digital television broadcast systems” describes the insertion of repetitive-PN1023 sequences into 8VSB digital television signals such that data segment sync signals are subsumed into the sequences. These longer sequences better accommodate the 90-microsecond or so possible range of echoes that was reported from field observations in spring 2000.
The channel impulse response (CIR) obtained in the response of a PN1023 match filter to a repetitive-PN1023 sequence inserted into an 8VSB digital television signal also provides a good basis for subsequent DFT computations of CIR under dynamic multipath reception conditions. These subsequent DFT computations involve sliding window procedures using auto-correlation of successive blocks of a few thousand consecutive 8VSB symbols throughout data fields. Such computations are described in U.S. Pat. No. 6,975,689 titled “Digital modulation signal receiver with adaptive channel equalization employing discrete Fourier transforms” granted on 13 Dec. 2005 to A. L. R. Limberg and J. D. McDonald. Further such computations are described in U.S. Pat. No. 7,050,491 titled “Adaptive equalization of digital modulating signal recovered from amplitude-modulated signal subject to multipath” granted on 23 May 2006 to J. D. McDonald and A. L. R. Limberg.
It was proposed that the ECR signal composed of repetitive-PN1023 sequences be put into data segments contiguous with the data segment containing the DFS signal. ATSC members deemed it highly desirable that an ECR signal not interfere with the operation of DTV signal receivers already in the field. Unfortunately, there were legacy DTV receivers already in the field that could not accommodate the cessation of trellis decoding other than for the initial segments of data fields containing DFS signals. So, the proposal to amend A/53 to permit insertion of ECR signals composed of repetitive-PN1023 sequences into DTV signals was not taken up.
In 2005 the DTV industry began to give serious attention to the problems of on-channel auxiliary transmitters for DTV signals used for implementing a single-frequency network (SFN). One of the problems was that there was no provision in A/53 for standardizing the phasing of the ⅔ trellis coding that converted bit pairs to bit triplets defining 8-level data symbols. Information to standardize the phasing of the ⅔ trellis coding for all transmitters broadcasting the same signal on the same channel was necessary. It was needed to avoid DTV receivers receiving the signal from more than one of the transmitters being unable to resolve phasing of the ⅔ trellis coding. Brief transition codes that could standardize phasing of the ⅔ trellis coding were known, but there remained the problem of how best to insert them into the fields of interleaved and trellis-coded data supplied to the DTV signal as modulating signal. The procedure used to synchronize ⅔ trellis coding (and pre-coding, if that were used to help overcome interference from co-channel NTSC signal) is called “determinate trellis resetting” or “DTR” for short.
Another problem associated with on-channel repeaters for DTV signals is that some DTV receivers receiving the signal from more than one transmitter will be subject to multipath reception afflicted with strong pre-echoes. These pre-echoes can be advanced by several microseconds and can be nearly as strong as the principal signal, which defines the “cursor” tap in time-domain equalization filtering. The equalization of DTV signals with strong pre-echoes is advantageously done by pre-filtering the signals so as to alter the time-domain spectrum to one in which the principal signal is substantially stronger than echo signals, especially pre-echo signals. The pre-filter can be one the kernel of which mirrors the time-domain channel impulse response (CIR) as described by R. W. Citta, S. M. LoPresto and J. Xia in U.S. Pat. No. 6,650,700 granted 18 Nov. 2003 and titled “Dual path ghost eliminating equalizer with optimum noise enhancement”. The pre-filter can be one which combines the received DTV signal with itself as subjected to suitable delay as A. L. R. Limberg described in his U.S. Pat. No. 7,151,797 granted 19 Dec. 2006 and titled “Adaptive K-factor-improvement filter for receiver of radio signals subject to multipath distortion”. A more complex pre-filter is described in U.S. Pat. No. 7,072,392 titled “Equalizer for Time Domain Signal Processing”, which issued 4 Jul. 2006 to J. Xia, R. W. Citta, S. M. LoPresto and W. Zhang.
The design of each of these pre-filters is predicated on knowledge of the CIR of the DTV signal as originally received. The need for expeditiously ascertaining CIR suggested that an ECR signal be inserted into the interleaved data field which ECR would support computation of CIR by a Dietrich-Greenberg type of method. The problem is that the ECR signal has to comply with the ⅔ trellis coding prescribed by A/53 in order not to disrupt the proper operation of legacy DTV receivers. The repetitive PN511 sequences described in U.S. Pat. No. 6,816,204 and the repetitive-PN1023 sequences described in U.S. Pat. No. 6,768,517 do not conform to symbols generated by ⅔ trellis coding because all the Z0 bits in the symbols are zero-valued. Approximating a repetitive PN sequence by symbols that could be generated by ⅔ trellis coding generally introduces Z0 bits that are variably ONEs and ZEROes. The presence of these variable Z0 bits compromises the unique auto-correlation properties of the repetitive PN sequence and introduces appreciable errors into the CIR generated by auto-correlation filtering.
The insertion of an ECR signal that occupies a number N of contiguous segments of the interleaved data field affects bytes in a number (N+51) data segments of the data field without interleaving. It would be desirable that these (N+51) data segments as so modified not disrupt the operation of legacy DTV receivers nor noticeably degrade reception.
In a U.S. published Pat. App. No. 20030021341 published 30 Jan. 2003 and entitled “Method of effective backwards compatible ATSC-DTV multipath equalization through training symbol induction” A. J. Vigil and M. A. Belkerdid describe ECR signals dispersed throughout the interleaved data field. Their patent application indicates that these ECR signals will compromise R-S codewords so as to interfere with the operation of legacy DTV receivers. In their patent application Vigil and Belkerdid suggested two mechanisms for enabling legacy DTV receivers to reject the compromised R-S codewords. One is to use a special packet identifier (PID) sequence in the headers of the codewords; the other is to modify the R-S codewords so legacy DTV receivers will find them to be incapable of being corrected. If an ECR signal is contiguous over N segments of the interleaved data field, it is probably impractical to use special PIDs to identify R-S codewords affected by bytes of ECR signal. U.S. published Pat. App. No. 20030021341 provides no specific description of how to insure that legacy DTV receivers will find modified R-S codewords to be incapable of being corrected.
U.S. Pat. No. 5,377,207 granted 27 Dec. 1994 to M. Perlman and titled “Mappings between codewords of two distinct (N, K) Reed-Solomon codes over GF (2J)”. Perlman points out that a Berlekamp alternative (N, K) R-S code is orthogonal to the conventional-architecture (N, K) R-S code. That is, R-S decoders in legacy receivers will find Berlekamp-architecture (207, 187) R-S codewords incapable of correction. If robust transmissions used Berlekamp-architecture (207, 187) R-S coding, apparently there would no longer be any need to worry about legacy receiver reception of ordinary 8VSB data being disrupted. Furthermore, substantially the same hardware can be used for decoding either type of (N, K) R-S code. According to U.S. Pat. No. 5,490,154 granted to R. Mester on 6 Feb. 1996 and titled “Method of and circuit arrangement for decoding RS-coded data signals”, Philips used substantially the same hardware for decoding either of the EBU D1 and ISO R-S codes employed in magnetic tape recording. It is known theoretically that other “orthogonal” (207, 187) R-S codes that sustain error correction besides the type described by Berlekamp exist. They differ from the A/53 standard (207, 187) R-S code in that their Galois fields are generated by primitive field generator polynomials with zero coefficient placements different from those in the primitive field generator polynomial shown in FIG. 6 of A/53 Annex D. (Other than in this paragraph, in this specification the variables N and K stand for things other than RS codeword length and data packet length.)